Electronic – 5.5 kHz noise on SMPS output

emi-filteringnoiseoscillationswitch-mode-power-supply

I am designing a low noise SMPS like this
http://www.ti.com/lit/df/tidrgc2/tidrgc2.pdf switching from 40-90 kHz.

I have a weird problem, on the output we are seeing a spike of noise at 5.5 kHz and multiple (decaying in a few multiples), tested with differential amplifier and correct techniques.

We have no idea where it comes from (maybe ringing of secondary transformer inductance + caps)?

Noise capture

Update:

We added a C and R across transformer, 1000 uF and 20 ohm and we seen a 6 dB reduction on noise. We are experimenting with different values to lower the noise. The only LC tank we are seeing is diode capacitance (and the RC snubber across it) and transformer secondary winding.

Second update.

We used 2x47uF parallel across pin 6/10 + 50R and spike is slightly lower .

Best Answer

Good question. People think that the only noise that can occur is the switching frequency and harmonics. This is not true. You are not the only one that has found noise below the switching frequency. Normal SM PS filter LC parts will work well at say 40 kHz but your 5.5 kHz will cut through it like butter. It will be better to nail this at the source than use a large expensive output filter. These limit cycle oscillations are sometimes associated with the resonant frequency of the output filter. Study this and study your control loop poles. Double check for interaction and change your loop or change your output filter. Loose ferrite cores can also cause audible oscillations despite running far above 20 kHz.

Related Solutions

Electronic – Why is SMPS controller IC failing

I have a suspicion that your problem might have to do with the bootstrap circuit, as detailed in section 5.3 of the data sheet. Some evidence leads to this conclusion:

You are likeley using larger MOSFETs than in the original design (because your overall power is greater), thus you are dealing with a higher gate charge, resulting in a higher current that must be delivered by the integrated bootstrap circuit. It could be that the bootstrap circuit dissipates more power than it was designed for and fails, sometimes resulting in a bad output driver for the high side MOSFET, sometimes taking the MOSFET with it. Adding an external, fast and high-voltage bootstrap diode might help. Note that the calculation in Eq. 9 of sect. 5.3 uses the typical on-resistance of the integrated bootstrap circuit. It's a better idea to use the max. value from table 4 which is twice as high.
Once the circuit goes into overload, you say that the frequency drops. During these prolonged on-times, the voltage across the external bootstrap capacitor might become too low to keep the high side MOSFET saturated, causing it to have a higher on-resistance, excessive losses and thermal overstress. However, in this case, it would be the high-side MOSFET that fails first. Check the voltage across the bootstrap capacitor. A bigger capacitor might help, but this will likely put more stress on the integrated bootstrap driver or the external diode that might be necessary anyway.

Another possibility could be that you are exceeding the maximum slew rates for the high-side driver. This might cause it to do weird things.

Concerning the negative spikes, an external protective clamping diode at each driver's output might help (K=Vout,HS; A=GND,LS and K=Vout,HS; A=GND,HS). Something as simple as a 1N4148 might be enough.

Electronic – ATX power supply front-end: How does this work

The schematic SmoothVTer included is based on an Asymmetric PWM Half Bridge Converter design. There are papers (search for "asymmetric half bridge converter") and documents (e.g. Fairchild's application notes) explaining the design details. Here are the specific answers to your questions:

Actually T1 is driven by the halved rectified DC voltage after the bridge rectifier BD1. You can see that if you remove the ZNR1 line altogether (the voltage select switch ZNR1 will be opened anyway when the input supply is 240V. I will explain the use of that below.) Therefore it is indeed the + and - DC output from BD1, halved by the pair of resistors R2 and R3 that is driving the power transformer.
The line with the ZNR1 switch going between R2/R3 C1/C2 allows the circuit to optionally operates as a simple rectifier (ZNR1 open) and/or Voltage Doubler (ZNR1 short.) See wiki's Voltage Doubler and/or Voltage Multiplier articles for more details. The varistor ZNR1 is often a physical switch for selecting between 240V (switch opened) and 110V (switch closed) in many ATX PSUs. (I am not sure if ZNR1 is really a varistor because that would make the switch open at 110V but short at 240V...) The two big caps (C1,C2) after the bridge rectifier BD1 is operating as a [Voltage Doubler][3] in the 110V (i.e. ZNR1 open) case. So the DC voltage across R2+R3 is always ~340V (240V*sqrt(2)) whether the input is 110V or 240V.
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How can any current flow in T1's primary at all? Cap C9 blocks all DC on T1's primary.
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Of course current can still flow in T1 with a capacitor, the voltage is being switch on and off all the time by PWM so it is effectively AC! Transformers relies on current/flux changes to operate. So cutting out the DC component is fine.
T2 is just operating as an isolation transformer (to keep the controller chip side away from dangerous high voltages) for the switching signals for operating the two NPN power transistors Q3 and Q4. The entire block between T2 and T1 (and including T1 actually) is the source side of the Asymmetric PWM Half Bridge Converter. The coil section between 10 and 5 is just used as the additional inductor connector in series needed for that circuit. The number of turns between 9 and 10 is the same as that between 6 and 7 but 5-10 is really a different inductor so 10 is generally not a centre tap.

Best Answer

Related Solutions

Electronic – Why is SMPS controller IC failing

Electronic – ATX power supply front-end: How does this work

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