I think you would want to simply use the circuit topology of Figure 5 of that Microchip App Note AN687. The circuit design appears to be pretty reasonable to me. With your lower range of signal you would need to increase the gain of A4 a bit. With your quoted range 80.3 to 175.8 mV you could raise the gain up to a value of about 13. Adjusting the gain setting resistors R10 and R11 to some standard values of R10=2.00K and R11=24.0K will result in this gain of 13. At a gain of 13 the output of A4 then will be in the range of 1.043V to 2.320V. With that simple change the design of the filter would not need to change at all.
If you did want to try to add in some offset to the signal so as to be able to use more of the A/D converter range at the low end then the filter design would likely have to change which adds more work. Although LT-Spice can be your friend in helping to analyze a filter design to see if you can get it to give the responses that you would expect.
Note that the whole App Note design is based on the premise that the opamps in the circuit are biased around a center point in the supplies of 0V. This means that you would need a bi-polar supply to the amplifier packages. For the low currents reqired in this circuit you should be able to easily produce a negative supply for your use. To do this I could recommend that you deploy the bog standard MAX232 chip as a voltage doubler and inverter to produce a nomimal -7 to -9 volts off the +5 volt supply. Then use a simple -5V linear regulator and a couple of capacitors to regulate the negative voltage down to -5V. A generic 7905 type part could do the trick here.
Actually, the chip you mention has something close to a current source- because they're measuring the voltage across the reference resistor they don't care if it changes a bit. Since the resistor (connected to the 2V bias) is 4x the 100°C value of the RTD and the RTD only changes by 30-40% for a +/-100°C range, the current is constant within about 10% at a (very high) 4mA for Pt100 and 0.4mA for a Pt1000. That is a much higher current than typically used in precision applications, so self heating is a definite source of error.
Let's take an example- you want to measure temperature from 0 to 100°C and you have a 0.8mA current source. Let's assume it's a Pt100 DIN curve (\$\alpha = 0.00385\$).
The voltage across the RTD will be 80mV at 0°C and 110.8mV at 100°C, for a span of 30.8mV. So a 0.1°C error (say that's our allowable error due to the electronics) would represent a 30.8uV voltage, and it would be 0.027% of full scale.
If you offset the 100 ohm base resistance of the sensor with a stable resistor (it's easy to get a resistor that is much more stable than a voltage or current source), and if we assume that error is relatively negligible, then we still have the 30.8uV error budget, but now we only have to have an accuracy of 0.1% in our measurement (almost 4 times better). A good ADC can be comparable to a precision resistor divider in ratiometric measurements, and that's what the MAX chip is depending on- also they're not shooting for the best possible accuracy, just something viable.
If you were thinking about using a circuit with, say, a 160mV voltage source and a series 100 ohm resistor to measure the current, you'd have substantially changing current through the sensor (so you'd get less resolution in degrees at high temperatures for a given resolution or noise floor), and the self-heating would greatly increase at low temperatures rather than appearing as a (relatively) fixed offset temperature. A high voltage V with a large series resistance R behaves the same as an imperfect current source of I = V/R with an output impedance equal to that R (Thevenin).
Best Answer
You shouldn't hook up multiple RTDs to a single current source in parallel. You would end up with more unknowns than equations. You measure the voltage across some particular RTD, but you don't know the excitation current into that particular RTD, and of course you don't know the resistance.
\$ \left\{ \begin{array}{l l} V_{RTD,1}=I_{exc,1}R_{RTD,1}\\ V_{RTD,2}=I_{exc,2}R_{RTD,2}\\ I_{exc} = I_{exc,2} + I_{exc,2} \end{array} \right. \$
I have left out the parasitic lead resistance.
\$I_{exc,1}\$, \$I_{exc,2}\$, \$R_{RTD,1}\$, \$R_{RTD,2}\$ are 4 unknowns. We have only 3 equations. So, this parallel connection doesn't work even for the case of 2x RTDs.
Either multiplex the current source and energize one RTD at a time. Or, provide an individual current source for each RTD.
If you hook up all of the RTDs in series, how are you going to measure and compensate for the lead in a 3-wire scheme? It should be possible, but I'm not sure that that would be simpler.
I would also be careful with your 2-channel A/D approach where a separate A/D conversion is made to cancel out the lead resistance in a 3-lead configuration. With an analog approach, you compensate for the lead instantly in analog . With 2-challel, you make a measurement of the RTD at a time τ1, and then you make another measurement of the lead at a time τ2. There will be 50 or 60 Hz noise from the mains. Due to noise, the state at τ2 may be different from τ1, and your compensation may introduce some error. You can alleviate this by filtering and/or canceling out the 50 or 60 Hz noise.
Analog Devices note CN-0287 describes a front end for measuring multiple RTDs. It uses a single current source. See p.4