oscilloscopeprobepspice
The oscilloscope and probe manufacturer gives me the following input impedance data:
Oscilloscope
Probe characteristics
What will be the PSPICE model
Like pretty much all real circuits, oscilloscope inputs have a parasitic capacitance. No matter how small you made it by good design, it would still affect RF signal acquisition, except maybe for a defined 50 Ω connection and attenuation directly at the scope's input, for which case, with the numbers from your question -
$$f_{-3dB} = \frac{1}{2\pi \cdot R_{in,\ scope} \cdot C_{in,\ scope}} = \frac{1}{2\pi \cdot 50 \;\Omega \cdot 12 \;pF} = 256 \;MHz $$
Or even higher, if we would make the scope's input impedance Cin, scope smaller.
Usually, though, we don't want to load the circuit under test with a defined 50 Ω connection because most circuits under test will have any impedance but 50 Ω (like your signal generator's output would, because it is specifically designed for impedance-matched 50 Ω systems). So what can be done with a capacitance that can't be eliminated? It was chosen to use it in a clever way in the probe-and-scope combination. So clever, actually, that any unknown capacitance that may be caused by probe cables and other things in your connection can be compensated just like the scope's input capacitance, and all of them become don't-cares for most cases of practical measurement applications.
The 1:10 probe has an internal resistor of 9 MΩ and, in parallel, an internal capacitor of [1/9 * Cin, scope].
It is adjustable because the probe doesn't know the exact capacitance of the particular scope it is connected to.
With the capacitor in the probe properly adjusted, you have not only a resistive divider for the DC part of the signal (9 MΩ at the probe vs. 1 MΩ in the scope), but also a capacitive divider for the higher-frequency AC part of the signal (1.33 pF at the probe vs. 12 pF in the scope, using your numbers), and the combination works beautifully up to or beyond, say, 500 MHz.
Also, you get the advantage of inserting not 1 MΩ and 12 pF into your circuit when probing, but 9 MΩ + 1 MΩ = 10 MΩ and [the series equivalent of 12 pF and (12 pF / 9)] = 1.2 pF
Link to the source of the picture: Here.
What the picture in the link doesn't show and what we have neglected so far is the capacitance of the probe's cable, this would just add to the capacitance at the scope's input and can also be compensated for when turning the variable cap in the probe.
Using a 1:10 probe, the probe's small capacitance is in series with the scope's larger input capacitance. The total capacitance (approx. 1.2 pF) is in parallel to the point of your circuit that you are probing. Connecting the scope directly to the circuit, e.g. with just a straight BNC cable, you are indeed putting the entire input capacitance of the scope in parallel to what you are measuring - maybe loading your circuit under test so much that it will not work any more while being measured. At best, it might still work somehow, but the picture on your scope will show results far off the real waveforms in your circuit under test.
It would be possible to build scopes with a much smaller input capacitance - but then, there would be no way compensating the probe's cable capacitance with a small variable capacitor near the probe tip. After all, the 12 pF at the scope's input have been put there on purpose, to make the scope work well together with a good probe.
One last note: Using 1:100 probes, you load your circuit even less. In lack of an active probe with a really small capacitance at the tip, a 1:100 probe can be used in cases where even 1.2 pF would be too much load on your circuit - provided the signal is large enough that you still see something after the probe's 1:100 attenuation.
Somethings not quite right there.
I have the 200MHz SD8202 which has the same manual so I assume the procedures are the same. It has a 5V pk-pk compensation output. I'm guessing you have adjusted the vertical position to zoom in at the top of the waveform.
I just tried this and I get similar to your top picture but with a bit less noise (did you connect the ground clip too? Was it bandwidth limited to 20MHz like the second waveform?)
All I can think of trying right now is running the self calibration routine (remove probes, press "Utility" button, press "Function" (H1 button), choose "Adjust" with "M" knob, press "Self Cal" (H2 button)
Let this complete (couple of minutes) and try the probe compensation again. Let us know how it goes.
EDIT - What do you get with all the waveform on the screen? (e.g. 1V/div)
It's possible the front end is slightly different in your model (or conditions/part tolerances are different), and it's at the limit of the vertical amplifier as Zebonaut proposes.
I'd also try a divider on the compensation signal to make it so the full waveform is on the screen at 20mV/div, just in case it's a problem specific to that gain setting.
At 5V pk-pk, to bring it down to say, 100mV pk-pk you need a 50:1 divider. Something like 50k and 1k will get "close enough" (98mV pk-pk) Or use a potentiometer and just adjust visibly.
If it still looks the same like this then I would say there is a problem.
Best Answer
It would look something like this:
A lot of this is assumed and depending on your requirements for accuracy, you will need to determine a lot of these values empirically for your probe. The values, especially for the compensation network will be highly specific to a given probe's rated bandwidth. Tip resistance can also be as high as 250Ω for lower bandwidth probes - regardless, it is definitely not zero. The AC coupled noise is added mostly for academic purposes. It will be so highly dependent on the environment, the probe's location and orientation within that environment, what lights are on, if someone is using an angle grinder within 100 feet, the angle of the tongue of an invisible cat that may or may not even exist, and all sorts of other things you can never hope to account for. As long as your probing techniques are decent, it should minimize the influence of any of this noise so for myself, I would just omit it. But it is shown for completeness' sake.
This is based entirely off of this excellent article in the October 2009 issue of Silicon Chip by Doug Ford which you can read here. I've summarized his final conclusions in the form of a SPICE, along with pertinent comments. In the article, transmission line parasitics of the cable are determined empirically and vary slightly depending on the bandwidth of the probe. Other values, such as those of the compensation network are determined largely by trial and error, by stepping through values and seeing which results in the minimal peaking and other effects, which is what the designers of the probe were also attempting to do.
Ultimately, you're going to have to determine a lot of stuff about this probe by measurement or just some trial and error where you try to make the simulation look like a real waveform capture of the same thing, and possibly disassembling the probe so you can measure components directly. Depending on how accurate or inaccurate you need the model to be, this can be either a fairly straight forward task with the understanding of a generous error margin being present, or a tedious and laborious task. Good luck.